1. Field of the Invention
The present invention relates to a highly efficient, small-sized, low-cost DC-DC converter.
2. Description of the Related Art
FIG. 1 is a circuit diagram of a single-output DC-DC converter that is a basic circuit for explaining a conventional multiple-output DC-DC converter.
The DC-DC converter shown in FIG. 1 is disclosed in Japanese Patent Application Publication No. 2003-319650, and is configured of a half-bridge circuit. In the circuit, the two sides of a DC power supply Vin are connected to a series circuit in which a switching element Q1 made of a metal oxide semiconductor field-effect transistor (MOSFET) and a switching element Q2 made of a MOSFET are connected in series. The drain terminal of the switching element Q2 is connected to the positive electrode of the DC power supply Vin, whereas the source terminal of the switching element Q1 is connected to the negative electrode of the DC power supply Vin.
Between the drain terminal and source terminal of the switching element Q1, a diode D1 and a voltage resonance capacitor Crv are connected in parallel, while a reactor Lr1, the primary winding P1 of a transformer T1, and a current resonance capacitor Cri are connected as a series circuit. The reactor Lr1 is configured of a leakage inductance between the primary winding and the secondary winding of the transformer T1. An exciting inductance, as a reactor Lp1, is equivalently connected to the primary winding P1. A diode D2 is connected in parallel between the drain terminal and source terminal of the switching element Q2.
A starting point of each winding of the transformer T1 is denoted by a dot (●). The anode of a diode D3 is connected to one end (● side) of the secondary winding S1 of the transformer T1. The other end of the secondary winding S1 of the transformer T1 and one end (● side) of a secondary winding S2 of the transformer T1 are connected to one end of a smoothing capacitor Co1. The other end of the secondary winding S2 of the transformer T1 is connected to the anode of a diode D4. The cathode of the diode D3 and the cathode of the diode D4 are connected to the other end of the capacitor Co1. A load Ro1 is connected to the both ends of the capacitor Co1.
On the basis of an output voltage Vo from the capacitor Co1, a control circuit 10 performs pulse-frequency-modulation (PFM) control (frequency control) by alternately turning on and off the switching elements Q1 and Q2 so that the output voltage V0 of the capacitor Co1 can be kept constant.
Detailed descriptions will be provided next for how a conventional DC-DC converter with the above configuration operates, with reference to a timing chart shown in FIG. 2.
In FIG. 2, VQ1 denotes the voltage between the drain terminal and source terminal of the switching element Q1; IQ1 denotes the drain terminal current of the switching element Q1; VQ2 denotes the voltage between the drain terminal and source terminal of the switching element Q2; IQ2 denotes the drain terminal current of the switching element Q2; VCri denotes the voltage between the two terminals of the current resonance capacitor Cri; VD3 denotes the voltage between the two terminals of the diode D3; ID3 denotes the current of the diode D3; VD4 denotes the voltage between the two terminals of the diode D4; and ID4 denotes the current of the diode D4.
It should be noted that: each of the switching elements Q1 and Q2 has a dead time for which the switching element is off; and the switching elements Q1 and Q2 alternately are turned on and off.
First of all, during time t0 to time t1, the switching element Q2 is turned from on to off at time t0. While the switching element Q2 is on, a current flows through Vin, Q2, Lr1, Lp1, Cri to Vin on the primary winding side of the transformer T1. A current flows through Co1, Ro1 to Co1 on the secondary winding side of the transformer T.
Once the switching element Q2 is turned off, the current which flows on the primary wiring side of the transformer T1 is commutated from the switching element Q2 to the voltage resonance capacitor Crv, and thus flows through Crv, Lr1, Lp1, Cri to Crv.
As a result, the voltage of the voltage resonance capacitor Crv is discharged down to zero volts, although being almost equal to the voltage of the direct voltage supply Vin while the switching element Q2 is turned on. Hereinafter, the voltage of the DC power supply Vin will also be denoted by the reference numeral Vin.
Accordingly, because the voltage of the voltage resonance capacitor Crv is equal to the voltage VQ1 of the switching element Q1, the voltage VQ1 of the switching element Q1 decreases from Vin to zero volts. In addition, because the voltage VQ2 of the switching element Q2 is expressed by (Vin−VQ1), the voltage VQ2 increases from zero volts to Vin.
During time t1 to time t2, once the voltage of the voltage resonance capacitor Crv decreases to zero volts at time t1, the diode D1 becomes conductive, and the current thus flows through D1, Lr1, Lp1 (P1), Cri to D1. In addition, the voltage of the secondary winding S2 of the transformer T1 reaches the output voltage Vo, and a current flows through Co1, Ro1 to Co1, whereas a current flows through S2, D4, Co1 to S2, on the secondary winding side of the transformer T1. Furthermore, when the gate terminal signal of the switching element Q1 is turned on during time t1 to time t2, the switching element Q1 is brought into a zero-voltage switching (ZVS) operation and a zero-current switching (ZCS) operation.
During time t2 to time t3, because the switching element Q1 is turned on at time t2, a current flows through Cri, Lp1 (P1), Lr1, Q1 to Cri, and the voltage Cri of the current resonance capacitor Cri decreases. In addition, on the secondary winding side of the transformer T1, a current flows through S2, D4, Co1 to S2, whereas a current also flows through Co1, Ro1 to Co1. The voltage of the secondary winding S2 is clamped by the voltage of the output voltage Vo, whereas the voltage of the primary winding P1 is clamped by a voltage obtained by multiplying the output voltage Vo by the turns ratio. As a result, a resonance current generated by the reactor Lr1 and the current resonance capacitor Cri, flows on the primary side of the transformer T1.
During time t3 to time t4, the voltage of the secondary winding S2 decreases below the output voltage Vo, and no current flows on the secondary winding side of the transformer T1. On the secondary winding side of the transformer T1, the current flows through Co1, Ro1 to Co1. In addition, on the primary winding side of the transformer T1, the current flows through Cri, Lp1, Lr1, Q1 to Cri, while on the primary winding side of the transformer T1, a resonance current generated by the sum (Lr1+Lp1) of the two reactors Lr1 and Lp1 as well as the current resonance capacitor Cri flows.
During time t4 to time t5, once the switching element Q1 is turned off at time t4, the current which has flown on the primary winding side of the transformer T1 is commutated from the switching element Q1 to the voltage resonance capacitor Crv, and a current flows through Lp1, Lr1, Crv, Lp1.
As a result, the voltage of the voltage resonance capacitor Crv, which has been almost equal to zero volts while the switching element Q1 is turned on, is charged up to Vin. Because the voltage of the voltage resonance capacitor Crv is equal to the voltage VQ1 of the switching element Q1, the voltage VQ1 of the switching element Q1 increases from zero volts to Vin. In addition, because the voltage VQ2 of the switching element Q2 is equal to (Vin−VQ1), the voltage VQ2 of the switching element Q2 decreases from Vin to zero volts.
During time t5 to time t6, once the voltage of the voltage resonance capacitor Crv increases up to Vin at time t5, the diode D2 becomes conductive, and a current flows through Lp1 (P1), Lr1, D2, Vin, Cri to Lp1 (P1). In addition, the voltage of the secondary winding S1 of the transformer T1 reaches the output voltage Vo, and the current flows through Co1, Ro1 to Co1, whereas a current flows through S1, D3, Co1 to S1, on the secondary winding side of the transformer T1. Furthermore, when the gate terminal signal of the switching element Q2 is turned on during the period from time t5 through time t6, the switching element Q2 is brought into a zero-voltage switching operation and a zero-current switching operation.
During time t6 to time t7, because the switching element Q2 is turned on at time t6, a current flows through Vin, Q2, Lr1, Lp1 (P1), Cri to Vin, and the voltage VCri of the current resonance capacitor Cri increases. In addition, a current flows through S1, D3, Co1 to S1, whereas the current flows through Co1, Ro1 to Co1, on the secondary winding side of the transformer T1. The voltage of the secondary winding S1 is clamped by the output voltage Vo, whereas the voltage of the primary winding P1 is clamped by a voltage obtained by multiplying the output voltage Vo by the turns ratio. As a result, the resonance current generated by the reactor Lr1 and the current resonance capacitor Cri, flows on the primary winding side of the transformer T1.
During time t7 to time t8, the voltage of the secondary winding S1 decreases below the output voltage Vo at time t7. The current flows through Co1, Ro1 to Co1. In addition, the current flows through Vin, Q2, Lr1, Lp1, Cri to Vin, on the primary winding side of the transformer T1, while the resonance current generated by the sum (Lr1+Lp1) of the two reactors Lr1 and Lp1 as well as the current resonance capacitor Cri, flows on the primary winding side of the transformer T1.
As described above, the conventional DC-DC converter shown in FIG. 1 controls the switching frequencies respectively of the switching elements Q1 and the switching element Q2 by use of the pulse signal with a duty of approximately 50%. Thereby, the conventional DC-DC converter changes the resonance current generated by the reactor Lr1, the reactor Lp1 and the current resonance capacitor Cri, so as to control the output voltage Vo. As a result, when the switching frequencies are increased, the output voltage Vo is decreased.
Furthermore, as shown in FIG. 1, a capacitor input system is adopted for the output smoothing means of this circuit. For this reason, if the secondary winding side of the transformer T1 is configured with multiple outputs, a multiple-output voltage supply circuit can be easily configured by: providing secondary windings S13, S14 in addition to the existing secondary windings S11 and S12 in the transformer T1a as shown in FIG. 3; and rectifying and smoothing a voltage generated in the secondary windings S13 and S14. In addition, because the secondary windings S11 and S12 as well as the secondary windings S13 and S14 are tightly coupled with one another, each of the multiple-output voltages with multiple outputs is in proportion to its turns ratio, resulting in good performance of cross regulation.
Because the output voltage Vo on the secondary winding side of the transformer T1 is in proportion to its turns ratio on the secondary winding side thereof as described above, the larger the number of windings on the secondary winding side of the transformer T1 is, the more finely the output voltage can be set.
In the case of the conventional circuit, however, the currents ID3 and ID4 flowing in the respective diodes D3 and D4 on the secondary winding side of the transformer 1 are each shaped like a sine curve because of the resonance current generated by the resonant capacitor Cri and the reactor Lr1. These sine curved currents flow, as ripple currents, into the smoothing capacitor Co1. As a result, when the output currents are large, large ripple currents also flow into the smoothing capacitor Co1. For example, if the two output voltages on the secondary winding side are 5 volts with 10 amperes and 24 volts with 2.1 amperes each with an output capacitance of 50 watts, then the capacitor with an output of 5 volts has approximately 5 times larger ripple current flowing thereinto than the capacitor with an output of 24 volts.
Moreover, when the number of turns is increased for each secondary winding, the number of turns also needs to be increased for the primary winding. As a result, the use of the leakage inductance between the primary winding and the secondary windings of the transformer T1 for the reactor Lr1 constituting the resonance circuit causes the following problem.
Specifically, when the number of turns of the primary winding of the transformer T1 is increased, the leakage inductance increases in proportion to the square of the its number of turns. In addition, the electric power transmitted to the secondary winding side of the transformer T1 is in proportion to the square root of Cri/Lr1, and the operating frequency is in proportion to Cri×Lr1.
As a result, if the operating frequency is constant, the output electric power is in proportion to the voltage Vri of the current resonance capacitor Cri. If a voltage supply circuit with a large output capacitance is configured, the current resonance capacitor Cri needs to be increased. If the operating frequency is constant, the reactor Lr1 needs to be decreased with the increase of current resonance capacitor Cri. In order to decrease the reactor Lr1, the number of turns of the primary winding needs to be decreased. If the voltage supply circuit with a large output electric power is configured, then the number of turns needs to be decreased, thereby making it difficult to accurately select the turns ratio on the secondary winding side.
To deal with this problem, in the conventional circuit configuration, a voltage supply with high efficiency and low noise can be configured by using zero-voltage switching (ZVS) and zero-current switching (ZCS). Moreover, in the same manner, a voltage supply circuit with a multiple-output voltage supply on the secondary winding side can be configured.
However, if the multiple-output voltage supply circuit with a large total electric power on the secondary winding side is configured by using the leakage inductance between the primary winding and the secondary winding of the transformer T1 for the reactor Lr1, then this configuration generates large ripple currents flowing into the smoothing capacitor Co1 because the currents flowing into the smoothing capacitor Co1 are shaped like sine curves. In addition, the large output of output currents produces heat and increases the ripple voltage, due to the ESR (equivalent series resistance) of the smoothing capacitor Co1. In order to solve this problem, the capacitance of the smoothing capacitor Co1 is increased, or the number of smoothing capacitor Co1 is increased.
Nevertheless, these solutions also have problems that the cost is increased, and that a reduction of the number of turns for the secondary windings does not allow the turns ratio to be set accurately. As a result, there is still a problem that any current solutions cannot deal with a voltage supply circuit with multiple outputs having lower output voltages.